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PDF NCP5424A Data sheet ( Hoja de datos )
| Número de pieza | NCP5424A | |
| Descripción | Dual Synchronous Buck Controller | |
| Fabricantes | ON Semiconductor | |
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NCP5424A
Dual Synchronous
Buck Controller with Input
Current Sharing
The NCP5424A is a flexible dual N−channel synchronous buck
controller utilizing V2™ control for fast transient response and
excellent line and load regulation. This highly versatile controller can
be configured as a single two phase output converter that draws
programmable amounts of current from two different input voltages or
all current from one supply. The NCP5424A can also be configured as
two independent out−of−phase controllers.
Using the NCP5424A in a current sharing input configuration is
ideal for applications where more power is required than is available
from one supply, such as video cards or other plug−in boards. When
configured as a dual output controller, the output of one controller can
be divided down and used as the reference for the second controller.
This tracking capability is useful in applications such as Double Data
Rate (DDR) Memory power where the termination voltage must track
VDD.
The NCP5424A provides a cycle−to−cycle current limit allowing
the system to handle transient overcurrent events. In addition, the
NCP5424A provides Soft Start, undervoltage lockout, and built−in
adaptive FET nonoverlap time to prevent shoot through.
Features
• Cycle−to−Cycle Current Limit
• Programmable Soft Start
• 100% Duty Cycle for Enhanced Transient Response
• 150 kHz to 600 kHz Programmable Frequency Operation
• Switching Frequency Set by Single Resistor
• Out−Of−Phase Synchronization Between the Channels Reduces the
Input Filter Requirement
• Undervoltage Lockout
Applications
• Video Graphics Card
• DDR Memory
• High Current (Two−Phase) Power Supplies
• Dual Output DC−DC Converters
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16
1
SO−16
D SUFFIX
CASE 751B
PIN CONNECTIONS AND
MARKING DIAGRAM
1
GATE(H)1
GATE(L)1
GND
BST
IS+1
IS−
VFB1
COMP1
16
GATE(H)2
GATE(L)2
VCC
ROSC
IS+2
VFB+2
VFB−2
COMP2
A = Assembly Location
WL = Wafer Lot
Y = Year
WW = Work Week
ORDERING INFORMATION
Device
NCP5424AD
NCP5424ADR2
Package
SO−16
SO−16
Shipping
48 Units/Rail
2500 Tape & Reel
© Semiconductor Components Industries, LLC, 2003
August, 2003 − Rev. 0
1
Publication Order Number:
NCP5424A/D
1 page  
NCP5424A
ELECTRICAL CHARACTERISTICS (continued) (0°C < TA < 70°C; 0°C < TJ < 125°C; ROSC = 30.9 k, CCOMP1,2 = 0.1 µF,
10.8 V < VCC < 13.2 V; 10.8 V < BST < 20 V, CGATE(H)1,2 = CGATE(L)1,2 = 1.0 nF, VFB+2 = 1.0 V; unless otherwise specified.)
Characteristic
Test Conditions
Min Typ Max
Oscillator
Switching Frequency
Switching Frequency
Switching Frequency
ROSC Voltage
Phase Difference
ROSC = 61.9 k; Measure GATE(H)1; Note 3
ROSC = 30.9 k; Measure GATE(H)1
ROSC = 15.1 k; Measure GATE(H)1; Note 3
ROSC = 30.9 k, Note 3
−
112
250
450
0.970
−
150
300
600
1.000
180
188
350
750
1.030
−
Supply Currents
VCC Current
BST Current
COMP1,2 = 0 V (No Switching)
COMP1,2 = 0 V (No Switching)
− 13 17
− 3.5 6.0
Undervoltage Lockout
Start Threshold
GATE(H) Switching; COMP1,2 charging
7.8 8.6 9.4
Stop Threshold
GATE(H) not switching; COMP1,2 discharging
7.0
7.8
8.6
Hysteresis
Start−Stop
0.5 0.8 1.5
Cycle−to−Cycle Current Limit
OVC Comparator Offset Voltage
0 V < IS+ 1, 2 < 5.5 V, 0 V < IS− < 5.5 V
55 70 85
IS+ 1, 2 Bias Current
0 V < IS+ 1, 2 < 5.5 V
−1.0 0.1
1.0
OVC Common Mode Range
− 0 − 5.5
OVC Latch COMP2 Discharge Current
COMP = 1.0 V
0.3 1.2 3.5
Discharge Threshold
− 0.20 0.25 0.30
IS− Bias Current
0 V < IS− < 5.5 V
−2.0 0.2
2.0
OVC Latch COMP1 Discharge Current
COMP1 = 1.0 V
2.0 5.0 8.0
3. Guaranteed by design, not 100% tested in production.
Unit
kHz
kHz
kHz
V
°
mA
mA
V
V
V
mV
µA
V
mA
V
µA
µA
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5
5 Page 
NCP5424A
DESIGN GUIDELINES
Definition of the design specifications
The output voltage tolerance can be affected by any or all
of the following:
1. buck regulator output voltage setpoint accuracy;
2. output voltage change due to discharging or charging
of the bulk decoupling capacitors during a load
current transient;
3. output voltage change due to the ESR and ESL of the
bulk and high frequency decoupling capacitors,
circuit traces, and vias;
4. output voltage ripple and noise.
Budgeting the tolerance is left to the designer who must
consider all of the above effects and provide an output
voltage that will meet the specified tolerance at the load.
The designer must also ensure that the regulator
component temperatures are kept within the manufacturer’s
specified ratings at full load and maximum ambient
temperature.
Selecting Feedback Divider Resistors
VOUT
R1
VFB
R2
Figure 7. Selecting Feedback Divider Resistors
The feedback pins (VFB1(2)) are connected to external
resistor dividers to set the output voltages. The error
amplifier is referenced to 1.0 V and the output voltage is
determined by selecting resistor divider values. Resistor R1
is selected based on a design trade−off between efficiency
and output voltage accuracy. The output voltage error can be
estimated due to the bias current of the error amplifier
neglecting resistor tolerance:
Error% + 1
10*6
1
R1 100%
R2 can be sized after R1 has been determined:
ǒ ǓR2 + R1
VOUT
1
*
1
Calculating Duty Cycle
The duty cycle of a buck converter (including parasitic
losses) is given by the formula:
Duty
Cycle
+
D
+
VOUT ) (VHFET ) VL)
VIN ) VLFET * VHFET *
VL
where:
VOUT = buck regulator output voltage;
VHFET = high side FET voltage drop due to RDS(ON);
VL = output inductor voltage drop due to inductor wire
DC resistance;
VIN = buck regulator input voltage;
VLFET = low side FET voltage drop due to RDS(ON).
Selecting the Switching Frequency
Selecting the switching frequency is a trade−off between
component size and power losses. Operation at higher
switching frequencies allows the use of smaller inductor and
capacitor values. Nevertheless, it is common to select lower
frequency operation because a higher frequency results in
lower efficiency due to MOSFET gate charge losses.
Additionally, the use of smaller inductors at higher
frequencies results in higher ripple current, higher output
voltage ripple, and lower efficiency at light load currents.
The value of the oscillator resistor is designed to be
linearly related to the switching period. If the designer
prefers not to use Figure 8 to select the necessary resistor, the
following equation quite accurately predicts the proper
resistance for room temperature conditions.
ROSC
+
21700 * fSW
2.31fSW
where:
ROSC = oscillator resistor in kΩ;
fSW = switching frequency in kHz.
800
700
600
500
400
300
200
100
10
20 30 40 50
ROSC (kW)
Figure 8. Switching Frequency
60
Selection of the Output Inductor
The inductor should be selected based on its inductance,
current capability, and DC resistance. Increasing the
inductor value will decrease output voltage ripple, but
degrade transient response. There are many factors to
consider in selecting the inductor including cost, efficiency,
EMI and ease of manufacture. The inductor must be able to
handle the peak current at the switching frequency without
saturating, and the copper resistance in the winding should
be kept as low as possible to minimize resistive power loss.
There are a variety of materials and types of magnetic
cores that could be used for this application. Among them
are ferrites, molypermalloy cores (MPP), amorphous and
powdered iron cores. Powdered iron cores are very
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| Páginas | Total 20 Páginas | |
| PDF Descargar | [ Datasheet NCP5424A.PDF ] | |
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