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PDF P6103ASU8 Data sheet ( Hoja de datos )

Número de pieza P6103ASU8
Descripción 5V/12V Synchronous-Rectified Buck Controller
Fabricantes uPI Semiconductor 
Logotipo uPI Semiconductor Logotipo



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uP6103
5V/12V Synchronous-Rectified
Buck Controller with Reference Input
General Description
Features
The uP6103 is a compact synchronous-rectified buck
controller specifically designed to operate from 5V or 12V
supply voltage and to deliver high quality output voltage as
low as 0.4V. These SOP-8 and PSOP-8 devices operate at
fixed 200/300 kHz frequency and provide an optimal level
of integration to reduce size and cost of the power supply.
The uP6103 supports both tracking mode and stand-alone
mode operation. The output voltage is tightly regulated to
the external reference voltage from 0.4V to 3.0V at tracking
mode or to internal 0.6V reference at stand-alone mode.
This controller integrates internal MOSFET drivers that
support 12V+12V bootstrapped voltage for high efficiency
power conversion. The bootstrap diode is built-in to simplify
the circuit design and minimize external part count.
Other features include internal softstart, undervoltage
protection, overcurrent protection and shutdown function.
With aforementioned functions, this part provides
customers a compact, high efficiency, well-protected and
cost-effective solutions. This part is available in SOP-8 and
PSOP-8 packages.
Ordering Information
Order Number Package Top Marking
Remark
uP6103S8 SOP-8L uP6103S8
200kHz
uP6103AS8 SOP-8L uP6103AS8
300kHz
uP6103ASU8 PSOP-8L uP6103ASU8
300kHz
Note: uPI products are compatible with the current IPC/
JEDEC J-STD-020 requirements. They are halogen-free,
RoHS compliant and 100% matte tin (Sn) plating that are
suitable for use in SnPb or Pb-free soldering processes.
† Operate from 5V or 12V Supply Voltage
† 3.3V to 12V VIN Input Range
† 0.6 VREF with 1.5% Accuracy
† Output Range from VREF to 80% of VIN
† Support Tracking Mode and Stand Alone Mode
Operation
† Simple Single-Loop Control Design
† Voltage-Mode PWM Control
† Fast Transient Response
† High-Bandwidth Error Amplifier
† 0% to 80% Duty Cycle
† Lossless, Programmable Overcurrent Protection
† Uses Lower MOSFET RDS(ON)
† 200/300 kHz Fixed Frequency Oscillator
† Internal Soft Start
† Integrated Bootstrap Diode
† RoHS Compliant and Halogen Free
Applications
† Power Supplies for Microprocessors or
Subsystem Power Supplies
† Cable Modems, Set Top Boxes, and xDSL
Modems
† Industrial Power Supplies; General Purpose
Supplies
† 5V or 12V Input DC-DC Regulators
† Low Voltage Distributed Power Supplies
Pin Configuration & Typical Application Circuit
BOOT
UGATE
GND
LGATE
18
27
36
45
SOP-8
PHASE
REFIN
FB
VCC
BOOT
UGATE
GND
LGATE
18
27
GND
36
45
PSOP-8
PHASE
REFIN
FB
VCC
Reference
Input
Disable
Enable
REFIN
7
VCC
5
BOOT
1
UGATE
2
8 PHASE
FB
6
Option
4 LGATE
3
GND
uPI Semiconductor Corp., http://www.upi-semi.com
Rev. F00, File Name: uP6103-DS-F0000
VIN
VOUT
1

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P6103ASU8 pdf
uP6103
Functional Description
to 5VDD in 66ms after the softstart cycle is intiated (for
uP6103). The ramp is created digitally, so there will be 100
small discrete steps. Accordingly, the output voltage will
follow the SS signal and ramp up smoothly to its target
level.
The SS signal keeps ramping up after it execeeds the
internal reference VREF. However, the reference voltage VREF
takes over the behavior of error amplifier after SS > VREF.
When the SS signal climb to its ceiling voltage (5V), the
uP6103 claims the end of softstart cyclce and enable the
under voltage protection of the output voltage.
For internal reference voltage, the effective ramp-up time of
the output voltage is about 3.6ms. For external reference
voltage, the effective ramp up time output voltage is
calculated as:
uP6103, 300kHz for uP6103A. Higher switching frequncy
allows higher control bandwidth and faster transient
response. However higher swithcing frequency results in
higher power loss in both controller and power MOSFETs.
The uP6103 detects PHASE voltage for the existence of
power input when the UGATE turns on the first time. If the
PHASE voltage does not exceed 3.0V when the UGATE
turns on, the uP6103 asserts that power input in not ready
and stops the softstart cycle. However, the internal SS
continues ramping up to 3.3VDD. Another softstart is
initiated after SS ramps up to 3.3VDD. The hiccup period
is about 6.4ms. Figure 3 shows the start up interval where
VIN does not present initially.
TSS = 6 × VREF
(ms)
Figure 2 shows a typical start up interval where the REFIN
pin has been released from a grounded (system shutdown)
state.
VIN
5V/Div
VOUT
5V/Div
VIN
5V/Div
VOUT
0 .5 V /D iv
LGATE
5V/Div
1ms/Div
LGATE
5 V /D iv
IX
2.5/Div
2ms/Div
Figure 2. Softstart Behavior.
Power Input Detection
The uP6103 detects PHASE voltage for the present of power
input when the UGATE turns on the first time. If the PHASE
voltage does not exceed 3.0V when the UGATE turns on,
the uP6103 asserts that power input in not ready and stops
the softstart cycle. However, the internal SS continues
ramping up to 5VDD. Another softstart is initiated after SS
ramps up to 5VDD. The hiccup period is about 12ms. Figure
3 shows the start up interval where VIN does not present
initially.
Switching Frequency
The switching frequency is fixed and can not be changed
externally. Typical switching frequency is 200kHz for
Figure 3. Softstart where VIN does not Present Initially.
Overcurrent Protection (OCP)
The uP6103 detects voltage drop across the lower MOSFET
(VPHASE) for overcurrent protection when it is turn on. If VPHASE
is lower than VOCP = -375mV, the uP6103 asserts OCP and
shuts down the converter.
Another factor should taken into consideration is the ripple
of the inductor current. The current near the valley of the
ripple current is used for OCP, resulting the averaged OCP
level a little higher than the calculated value.
Undervoltage Protection (UVP)
The FB voltage is monitored for undervoltage protection.
The UVP threshold level is typical 0.4V for both stand-
alone and tracking mode. The uP6103 shuts down upon
the detection of UVP and can be reset only by POR or
toggling REFIN pin.
uPI Semiconductor Corp., http://www.upi-semi.com
Rev. F00, File Name: uP6103-DS-F0000
5

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P6103ASU8 arduino
uP6103
Application Information
Power MOSFET Selection
External component selection is primarily determined by
the maximum load current and begins with the selection of
power MOSFET switches. The uP6103 requires two
external N-channel power MOSFETs for upper (controlled)
and lower (synchronous) switches. Important parameters
for the power MOSFETs are the breakdown voltage V(BR)DSS,
on-resistance RDS(ON), reverse transfer capacitance CRSS,
maximum current IDS(MAX), gate supply requirements, and
thermal management requirements.
The gate drive voltage is powered by VCC pin that receives
4.5V~13.2V supply voltage. When operating with a 12V
power supply for VCC (or down to a minimum supply
voltage of 8V), a wide variety of NMOSFETs can be used.
Logic-level threshold MOSFET should be used if the input
voltage is expected to drop below 8V. Since the lower
MOSFET is used as the current sensing element, particular
attention must be paid to its on-resistance. Look for RDS(ON)
ratings at lowest gate driving voltage.
Special cautions should be exercised on the lower switch
exhibiting very low threshold voltage VGS(TH). The shoot-
through protection present aboard the uP6103 may be
circumvented by these MOSFETs if they have large
parasitic impedences and/or capacitances that would inhibit
the gate of the MOSFET from being discharged below its
threshold level before the complementary MOSFET is
turned on. Also avoid MOSFETs with excessive switching
times; the circuitry is expecting transitions to occur in under
50 nsec or so.
In high-current applications, the MOSFET power
dissipation, package selection and heatsink are the
dominant design factors. The power dissipation includes
two loss components; conduction loss and switching loss.
The conduction losses are the largest component of power
dissipation for both the upper and the lower MOSFETs.
These losses are distributed between the two MOSFETs
according to duty cycle. Since the uP6103 is operating in
continuous conduction mode, the duty cycles for the
MOSFETs are:
DUP
=
VOUT
VIN
;
DLO
=
VIN
VOUT
VIN
The resulting power dissipation in the MOSFETs at
maximum output current are:
PUP = IO2 UT × RDS(ON) × DUP + 0.5 × IOUT × VIN × TSW × fOSC
PLO = IO2 UT × RDS(ON) × DLO
where TSW is the combined switch ON and OFF time.
Both MOSFETs have I2R losses and the top MOSFET
includes an additional term for switching losses, which are
largest at high input voltages. The bottom MOSFET losses
are greatest when the bottom duty cycle is near 100%,
during a short-circuit or at high input voltage. These
equations assume linear voltage current transitions and do
not adequately model power loss due the reverse-recovery
of the lower MOSFET’s body diode. Ensure that both
MOSFETs are within their maximum junction temperature
at high ambient temperature by calculating the temperature
rise according to package thermal-resistance
specifications. A separate heatsink may be necessary
depending upon MOSFET power, package type, ambient
temperature and air flow.
The gate-charge losses are dissipated by the uP6103 and
don’t heat the MOSFETs. However, large gate charge
increases the switching interval, TSW that increases the
MOSFET switching losses. The gate-charge losses are
calculated as:
PG = VCC × (VCC × (CISS _ UP + CISS _ LO ) + VIN × CRSS ) × fOSC
where CISS_UP is the input capacitance of the upper
MOSFET, CISS_LO is the input capacitance of the lower
MOSFET, and CRSS_UP is the reverse transfer capacitance
of the upper MOSFET. Make sure that the gate-charge loss
will not cause over temperature at uP6103, especially with
large gate capacitance and high supply voltage.
Output Inductor Selection
Output inductor selection usually is based the
considerations of inductance, rated current, size
requirement, and DC resistance (DC)
Given the desired input and output voltages, the inductor
value and operating frequency determine the ripple current:
IL
=
1
fOSC × LOUT
× VOUT
× (1
VOUT
VIN
)
Lower ripple current reduces core losses in the inductor,
ESR losses in the output capacitors and output voltage
ripple. Highest efficiency operation is obtained at low
frequency with small ripple current. However, achieving this
requires a large inductor. There is a tradeoff between
component size, efficiency and operating frequency. A
reasonable starting point is to choose a ripple current that
is about 40% of IOUT(MAX).
There is another tradeoff between output ripple current/
voltage and response time to a transient load. Increasing
the value of inductance reduces the output ripple current
and voltage. However, the large inductance values reduce
the converter’s response time to a load transient.
uPI Semiconductor Corp., http://www.upi-semi.com
Rev. F00, File Name: uP6103-DS-F0000
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