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PDF NCV8852 Data sheet ( Hoja de datos )
| Número de pieza | NCV8852 | |
| Descripción | Automotive Grade Non-Synchronous Buck Controller | |
| Fabricantes | ON Semiconductor | |
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NCV8852
Automotive Grade
Non-Synchronous Buck
Controller
The NCV8852 is an adjustable−output non−synchronous buck
controller which drives an external P−channel MOSFET. The device
uses peak current mode control with internal slope compensation. The
IC incorporates an internal regulator that supplies charge to the gate
driver.
Protection features include internal soft−start, undervoltage lockout,
cycle−by−cycle current limit, hiccup−mode overcurrent protection,
hiccup−mode short−circuit protection.
Additional features include: programmable switching frequency,
low quiescent current sleep mode and externally synchronizable
switching frequency.
Features
• Ultra Low Iq Sleep Mode
• Adjustable Output with 800 mV ±2.0% Reference Voltage
• Wide Input of 3.1 to 44 V with Undervoltage Lockout (UVLO)
• Programmable Switching Frequency
• Internal Soft−Start (SS)
• Fixed−Frequency Peak Current Mode Control
• Internal Slope Compensating Artificial Ramp
• Internal High−Side PMOS Gate Driver
• Regulated Gate Driver Current Source
• External Frequency Synchronization (SYNC)
• Programmable Cycle−by−Cycle Current Limit (CL)
• Hiccup Overcurrent Protection (OCP)
• Output Short Circuit Hiccup Protection (SCP)
• Space−Saving 8−PIN SOIC Package
• NCV Prefix for Automotive and Other Applications Requiring
Unique Site and Control Change Requirements; AEC−Q100
Qualified and PPAP Capable
• These Devices are Pb−Free and are RoHS Compliant
www.onsemi.com
8
1
SOIC−8
SUFFIX D
CASE 751
MARKING
DIAGRAM
8
V8852xx
ALYW
G
1
V8852xx= Specific Device Code
xx = (blank), 01
A = Assembly Location
L = Wafer Lot
Y = Year
W = Work Week
G = Pb−Free Package
PINOUT DIAGRAM
1 ROSC
2 EN/SYNC
3 COMP
4 FB
VIN 8
ISNS 7
GDRV 6
GND 5
ORDERING INFORMATION
Device
Package
Shipping†
NCV8852DR2G
SOIC−8 2500/Tape & Reel
(Pb−Free)
NCV885201D1R2G SOIC−8 2500/Tape & Reel
(Pb−Free)
†For information on tape and reel specifications,
including part orientation and tape sizes, please
refer to our Tape and Reel Packaging Specification
Brochure, BRD8011/D.
© Semiconductor Components Industries, LLC, 2015
August, 2015 − Rev. 7
1
Publication Order Number:
NCV8852/D
1 page  
NCV8852
ELECTRICAL CHARACTERISTICS
(VIN = 3.4 V to 36 V, EN = 5 V. Min/Max values are valid for the temperature range −40°C ≤ TJ ≤ 150°C unless noted otherwise, and are
guaranteed by test, design or statistical correlation)
Characteristic
Symbol
Conditions
Min Typ Max Unit
CURRENT SENSE AMP
Input Bias Current
Isns,bias
NCV8852
NCV885201
30 50 mA
70 120
CURRENT LIMIT / OVER CURRENT PROTECTION
Cycle−by−Cycle Current Limit
Threshold
Vcl
85 100 115 mV
Cycle−by−Cycle Current Limit
Response Time
tcl
200 nsec
Over Current Protection
Threshold
Vocp
% of Vcl
125 150 175
%
Over Current Protection
Response Time
tocp
200 ns
GATE DRIVERS
Leading Edge Blanking Time
Gate Driver Pull Up Current
Gate Driver Pull Down
Current
ton,min
Isink
Isrc
VIN − VGDRV = 4 V
VIN − VGDRV = 4 V
100 ns
200 300 mA
200 300 mA
Gate Driver Clamp Voltage
(VIN – VGDRV)
Vdrv
Power Switch Gate to Source Vgs VIN = 4 V
Voltage
6.0 8.0 10
3.8
V
V
SHORT CIRCUIT PROTECTION
Startup Blanking Time
Short−Circuit Threshold
Voltage
tscp,dly
Vscp
From start of soft−start, % of soft−start time
% of Feedback Voltage (Vref)
105 300 %
65 70 75 %
Hiccup Time
SC Response Time
THERMAL SHUTDOWN
Thermal Shutdown Threshold
Thermal Shutdown
Hysteresis
thcp,dly
tscp
Tsd
Tsd,hys
% of Soft−Start Time
Switcher Running
TJ rising
TJ Shutdown – TJ Startup
135
60 200
%
ns
160 170 180
10 15 20
°C
°C
Thermal Shutdown Delay
ttsd TJ > Thermal Shutdown Threshold to stop
switching
200 ns
www.onsemi.com
5
5 Page 
NCV8852
(4) MOSFET Selection
The NCV8852 has been designed to work with a
P−channel MOSFET in a non−synchronous buck
configuration. The MOSFET needs to be capable of
handling the maximum allowable current in the system, ICL.
Keep in mind that, depending on your minimum VIN signal,
it is possible to achieve 100% duty cycle. The power
dissipated through the MOSFET during conduction is as
follows:
PMOS,on + ICL 2 @ DMAX @ rDS,on
where:
PMOS,on: power through MOSFET [W]
ICL: cycle−by−cycle current limit [A]
rDS,on: on−resistance of the MOSFET [W]
To calculate the switching losses through the MOSFET, use
the following equation:
PMOS,sw
+
1
2
VIN
@
IOUT
@
ǒton
)
toffǓ
@
FSW
ton
+
toff
+
QGate
Idrv
where:
PMOS, sw: MOSFET switching losses [W]
ton: time to turn on the MOSFET [s]
toff: time to turn off the MOSFET [s]
QGate: gate charge [C]
Idrv: gate drive current [A]
(5) Diode Selection
The diode must be chosen according to its maximum
current and voltage ratings, and to thermal considerations.
The maximum reverse voltage the diode sees is the
maximum input voltage (with some margin in case of
ringing on the switch node). The maximum forward current
is the peak current limit of the NCV8852, or 150% of ICL.
(6) Output Inductor Selection
Both mechanical and electrical considerations influence
the selection of an output inductor. From a mechanical
perspective, smaller inductor values generally correspond to
smaller physical size. Since the inductor is often one of the
largest components in the power supply, a minimum
inductor value is particularly important in space−
constrained applications. From an electrical perspective, an
inductor is chosen for a set amount of current ripple and to
assure adequate transient response.
The output inductor controls the current ripple that occurs
over a switching period. A high current ripple will result in
excessive power loss and ripple current requirements. A low
current ripple will result in a poor control signal and a slow
current slew rate in the event of a load transient. A good
starting point for peak−to−peak ripple is around 10% of the
inductor current.To choose the inductor value based on the
peak−to−peak ripple current, use the following equation:
iL
+
VOUT @ (1 * DMIN)
L @ FSW
where:
iL: peak−to−peak output current ripple [Ap−p]
From this equation it is clear that the ripple current increases
as L decreases, emphasizing the trade−off between dynamic
response and ripple current. The peak and valley values of
the triangular current waveform are as follows:
IL(pk)
+
IOUT
)
iL
2
IL(vly)
+
IOUT
*
iL
2
where:
IL(pk): peak (maximum) value of ripple current [A]
IL(vly): valley (minimum) value of ripple current [A]
Saturation current is specified by inductor manufacturers as
the current at which the inductance value has dropped from
the nominal value, typically 10%. For stable operation, the
output inductor must be chosen so that the inductance is
close to the nominal value even at the peak output current,
IL(pk), it is recommended to choose an inductor with
saturation current sufficiently higher than the peak output
current, such that the inductance is very close to the nominal
value at the peak output current. This allows for a safety
factor and allows for more optimized compensation.
Inductor efficiency is another consideration when
selecting an output inductor. Inductor losses include dc and
ac winding losses, which are very low due to high core
resistance, and magnetic hysteresis losses, which increase
with peak−to−peak ripple current. Core losses also increase
as switching frequency increases.
Ac winding losses are based on the ac resistance of the
winding and the RMS ripple current through the inductor,
which is much lower than the dc current. The ac winding
losses are due to skin and proximity effects and are typically
much less than dc losses, but increase with frequency. Dc
winding losses account for a large percentage of output
inductor losses and are the dominant factor at switching
frequencies at or below 500 kHz. The dc winding losses in
the inductor can be calculated with the following equation:
PL(dc) + IOUT 2 @ Rdc
where:
PL(dc): dc winding losses in the output inductor
Rdc: dc resistance of the output inductor (DCR)
www.onsemi.com
11
11 Page | ||
| Páginas | Total 13 Páginas | |
| PDF Descargar | [ Datasheet NCV8852.PDF ] | |
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