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PDF LM20343 Data sheet ( Hoja de datos )
| Número de pieza | LM20343 | |
| Descripción | 3A Adjustable Frequency Synchronous Buck Regulator | |
| Fabricantes | National Semiconductor | |
| Logotipo |  |
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www.DataSheet4U.com
August 12, 2008
LM20343
36V, 3A Adjustable Frequency Synchronous Buck
Regulator
General Description
The LM20343 is a full featured synchronous buck regulator
capable of delivering up to 3A of load current. The current
mode control loop is externally compensated with only two
components, offering both high performance and ease of use.
The device is optimized to work over the input voltage range
of 4.5V to 36V making it well suited for high voltage systems.
The device features internal Over Voltage Protection (OVP)
and Over Current Protection (OCP) circuits for increased sys-
tem reliability. A precision Enable pin and integrated UVLO
allows the turn on of the device to be tightly controlled and
sequenced. Startup inrush currents are limited by both an in-
ternally fixed and externally adjustable soft-start circuit. Fault
detection and supply sequencing are possible with the inte-
grated power good (PGOOD) circuit.
The frequency of this device can be adjusted from 250 kHz to
1.0 MHz by connecting an external resistor from the RT pin
to ground.
The LM20343 is designed to work well in multi-rail power
supply architectures. The output voltage of the device can be
configured to track a higher voltage rail using the SS/TRK pin.
If the output of the LM20343 is pre-biased at startup it will not
sink current to pull the output low until the internal soft-start
ramp exceeds the voltage at the feedback pin.
The LM20343 is offered in an exposed pad 20-pin eTSSOP
package that can be soldered to the PCB, eliminating the
need for bulky heatsinks.
Features
■ 4.5V to 36V input voltage range
■ 3A output current, 5.2A peak current
■ 130 mΩ/110 mΩ integrated power MOSFETs
■ 94% peak efficiency with synchronous rectification
■ 1.5% feedback voltage accuracy
■ Current mode control, selectable compensation
■ Resistor programmed, 1MHz capable oscillator
■ Adjustable output voltage down to 0.8V
■ Compatible with pre-biased loads
■ Programmable soft-start with external capacitor
■ Precision enable pin with hysteresis
■ OVP, UVLO inputs and PGOOD output
■ Internally protected with peak current limit, thermal
shutdown and restart
■ Accurate current limit minimizes inductor size
■ Non-linear current mode slope compensation
■ eTSSOP-20 exposed pad package
Applications
■ Simple to design, high efficiency point of load regulation
from a 4.5V to 36V bus
■ High Performance DSPs, FPGAs, ASICs and
Microprocessors
■ Communications Infrastructure, Automotive
Simplified Application Circuit
© 2008 National Semiconductor Corporation 300517
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1 page  
Efficiency vs. Load Current
www.DataSheet4U.com fSW = 750 kHz
Error Amplifier Gain
Error Amplifier Phase
30051785
Line Regulation
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Load Regulation
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VCC vs. VIN
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5
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prevent catastrophic failures from accidental device over-
wwhwea.Dtinagta.Sheet4U.com
Design Guide
This section walks the designer through the steps necessary
to select the external components to build a fully functional
power supply. As with any DC-DC converter numerous trade-
offs are possible to optimize the design for efficiency, size, or
performance. These will be taken into account and highlight-
ed throughout this discussion. To facilitate component selec-
tion discussions the circuit shown in Figure 1 below may be
used as a reference. Unless otherwise indicated all formulas
assume units of amps (A) for current, farads (F) for capaci-
tance, henries (H) for inductance and volts (V) for voltages.
FIGURE 1. Typical Application Circuit
30051729
The first equation to calculate for any buck converter is duty-
cycle. Ignoring conduction losses associated with the FETs
and parasitic resistances it can be approximated by:
INDUCTOR SELECTION (L)
The inductor value is determined based on the operating fre-
quency, load current, ripple current and duty cycle.
The inductor selected should have a saturation current rating
greater than the peak current limit of the device. Keep in mind
the specified current limit does not account for delay of the
current limit comparator, therefore the current limit in the ap-
plication may be higher than the specified value. To optimize
the performance and prevent the device from entering current
limit at maximum load, the inductance is typically selected
such that the ripple current, ΔiL, is not greater than 30% of the
rated output current. Figure 2 illustrates the switch and in-
ductor ripple current waveforms. Once the input voltage, out-
put voltage, operating frequency and desired ripple current
are known, the minimum value for the inductor can be calcu-
lated by the formula shown below:
30051767
FIGURE 2. Switch and Inductor Current Waveforms
If needed, slightly smaller value inductors can be used, how-
ever, the peak inductor current, IOUT + ΔiL/2, should be kept
below the peak current limit of the device. In general, the in-
ductor ripple current, ΔiL, should be more than 10% of the
rated output current to provide adequate current sense infor-
mation for the current mode control loop. If the ripple current
in the inductor is too low, the control loop will not have suffi-
cient current sense information and can be prone to instability.
OUTPUT CAPACITOR SELECTION (COUT)
The output capacitor, COUT, filters the inductor ripple current
and provides a source of charge for transient load conditions.
A wide range of output capacitors may be used with the
LM20343 that provide excellent performance. The best per-
formance is typically obtained using ceramic, SP or OSCON
type chemistries. Typical trade-offs are that the ceramic ca-
pacitor provides extremely low ESR to reduce the output
ripple voltage and noise spikes, while the SP and OSCON
capacitors provide a large bulk capacitance in a small volume
for transient loading conditions.
When selecting the value for the output capacitor, the two
performance characteristics to consider are the output volt-
age ripple and transient response. The output voltage ripple
can be approximated by using the following formula:
where, ΔVOUT (V) is the amount of peak to peak voltage ripple
at the power supply output, RESR (Ω) is the series resistance
of the output capacitor, fSW(Hz) is the switching frequency,
and COUT (F) is the output capacitance used in the design.
The amount of output ripple that can be tolerated is applica-
tion specific; however a general recommendation is to keep
the output ripple less than 1% of the rated output voltage.
Keep in mind ceramic capacitors are sometimes preferred
because they have very low ESR; however, depending on
package and voltage rating of the capacitor the value of the
capacitance can drop significantly with applied voltage. The
output capacitor selection will also affect the output voltage
droop during a load transient. The peak droop on the output
voltage during a load transient is dependent on many factors;
however, an approximation of the transient droop ignoring
loop bandwidth can be obtained using the following equation:
11 www.national.com
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| Páginas | Total 20 Páginas | |
| PDF Descargar | [ Datasheet LM20343.PDF ] | |
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