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Número de pieza LM2647LQX
Descripción Dual Synchronous Buck Regulator Controller
Fabricantes National Semiconductor 
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June 2003
LM2647
Dual Synchronous Buck Regulator Controller
General Description
The LM2647 is an adjustable 200-500kHz dual channel
voltage-mode controlled high-speed synchronous buck
regulator controller ideally suited for battery powered appli-
cations such as laptop and notebook computers. The
LM2647 requires only N-channel FETs for both the upper
and lower positions of each synchronous stage. It features
line feedforward to improve the response to input transients.
At very light loads, the user can choose between the high-
efficiency Pulse-skip mode or the constant frequency
Forced-PWM mode. Lossless current limiting without the use
of external sense resistors is made possible by sensing the
voltage drop across the bottom FET. A unique adaptive duty
cycle clamping technique is incorporated to significantly re-
duce peak currents under abnormal load conditions. The two
independently programmable outputs switch 180˚ out of
phase (interleaved switching) to reduce the input capacitor
and filter requirements. The input voltage range is 5.5V to
28V while the output voltages are adjustable down to 0.6V.
Standard supervisory and control features include Soft-start,
Power Good, output Under-voltage and Over-voltage protec-
tion, Under-voltage Lockout, Soft-shutdown and Enable.
Features
n Input voltage range from 5.5V to 28V
n Synchronous dual-channel Interleaved switching
n Forced-PWM or Pulse-skip modes
n Lossless bottom-side FET current sensing
n Adaptive duty cycle clamping
n High current N-channel FET drivers
n Low shutdown supply currents
n Reference voltage accurate to within ±1.5%
n Output voltage adjustable down to 0.6V
n Power Good flag and Chip Enable
n Under-voltage lockout
n Over-voltage/Under-voltage protection
n Soft-start and Soft-shutdown
n Switching frequency adjustable 200kHz-500kHz
Applications
n Notebook Chipset Power Supplies
n Low Output Voltage High-Efficiency Buck Regulators
Typical Application (Channel 2 in parenthesis)
© 2003 National Semiconductor Corporation DS200563
20056304
www.national.com

1 page




LM2647LQX pdf
Absolute Maximum Ratings (Note 1)
If Military/Aerospace specified devices are required,
please contact the National Semiconductor Sales Office/
Distributors for availability and specifications.
Voltages from the indicated pins to SGND/PGND unless
otherwise indicated (Note 2):
VIN 30V
V5 7V
VDD
7V
BOOT1, BOOT2
36V
BOOT1 to SW1, BOOT2 to
SW2
7V
SW1, SW2
30V
ILIM1, ILIM2
30V
SENSE1, SENSE2, FB1, FB2
7V
PGOOD
7V
EN 7V
Power Dissipation (TA = 25˚C)
(Note 3)
Junction Temperature
ESD Rating (Note 4)
Ambient Storage Temperature
Range
Soldering Dwell Time,
Temperature
Wave
Infrared
Vapor Phase
1.0W
+150˚C
2kV
-65˚C to +150˚C
4 sec, 260˚C
10 sec, 240˚C
75 sec, 219˚C
Operating Ratings (Note 1)
VIN
VDD, V5
Junction Temperature
5.5V to 28V
4.5V to 5.5V
-5˚C to +125˚C
Electrical Characteristics
Specifications with standard typeface are for TJ = 25˚C, and those with boldface apply over full Operating Junction Tempera-
ture range. VDD = V5 = 5V, VSGND = VPGND = 0V, VIN = 15V, VEN = 3V, RFADJ = 22.1K unless otherwise stated. (Note 5)
Symbol
Parameter
Conditions
Min Typical Max
(Note 6) (Note 7) (Note 6)
Units
Reference
VFB_REG
FB Pin Voltage at Regualtion VDD = 4.5V to 5.5V,
(either FB Pin)
VIN = 5.5V to 28V
591 600 609 mV
VFB Line Regulation
VDD = 4.5V to 5.5V,
VIN = 5.5V to 28V
0.5
IFB FB Pin Current (sourcing)
Chip Supply
VFB at regulation
20 100 nA
IQ_VIN
ISD_VN
IQ_VDD
ISD_VDD
IQ_V5
ISD_V5
IQ_BOOT
ISD_BOOT
VUVLO
Logic
VIN Quiescent Current
VIN Shutdown Current
VDD Quiescent Current
VDD Shutdown Current
V5 Normal Operating Current
V5 Shutdown Current
BOOT Quiescent Current
BOOT Shutdown Current
VDD UVLO Threshold
VDD UVLO Hysteresis
VFB1 = VFB2 = 0.7V
VEN = 0V
VFB1 = VFB2 = 0.7V
VEN = 0V
VFB1 = VFB2 = 0.7V
VFB1 = VFB2 = 0.5V
VEN = 0V
VFB1 = VFB2 = 0.7V
VFB1 = VFB2 = 0.5V
VEN = 0V
VDD rising from 0V
VDD = V5 falling from VUVLO
100 200
µA
0 5 µA
2.5 4 mA
8 15 µA
0.3 0.5 mA
1 1.5
0 5 µA
2 5 µA
300 500
1 5 µA
3.9 4.2 4.5
V
0.5 0.7 0.9
V
IEN
VEN_HI
VEN_LO
VFPWM_HI
VFPWM_LO
EN Input Current
EN Input Logic High
EN Input Logic Low
FPWM Pull-down
FPWM Input Logic High
FPWM Input Logic Low
VEN = 0 to 5V
VFPWM = 2V
0 µA
2 1.8
V
1.3 0.8
V
100 200 1000 k
2 1.8
V
1.3 0.8
V
5 www.national.com

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LM2647LQX arduino
Operation Descriptions (Continued)
In a conventional converter, as the load is decreased to
about 10-30% of maximum load current, DCM (Discontinu-
ous Conduction Mode) occurs. In this condition the inductor
current falls to zero during the OFF-time, and stays there
until the start of the next switching cycle. In this mode, if the
load is decreased further, the duty cycle decreases (pinches
off), and ultimately may decrease to the point where the
required pulse width becomes less than the minimum ON-
time achievable by the converter (controller + FETs). Then a
sort of random skipping behavior occurs as the error ampli-
fier struggles to maintain regulation. This is not the most
desirable type of behavior. There are two ways out of this
problem.
One way is to keep the lower FET ON until the start of the
next cycle (as in the LM2647 operated in FPWM mode). This
allows the inductor current to drop to zero and then actually
reverse direction (negative direction through inductor, pass-
ing from Drain to Source of lower FET, see Channel 4 in
Figure 2). Now the current can continue to flow continuously
till the end of the switching cycle. This maintains CCM and
so the duty cycle does not start to pinch off as in typical
DCM. Nor does it lead to the undesirable random skipping
described above. Note that the pulse width (duty cycle) for
CCM is virtually constant for any load and therefore does not
usually run into the minimum ON-time restriction. But it can
happen, especially when the application consists of a very
high input voltage, a low output voltage rail, and also the
switching frequency is set high. Let us check the LM2647 to
rule out this remote possibility. For example, with an input of
24V, an output of 1V, the duty cycle is 1/24 = 4.2%. This
leads to a required ON-time of 0.042* 3.3 = 0.14 µs at a
switching frequency of 300kHz (T=3.3 µs). Since 140ns
exceeds the minimum ON-time of 30ns of the LM2647,
normal constant frequency CCM mode of operation is as-
sured in FPWM mode, at virtually any load.
The second way out of the problems of discontinuous mode
is the second operating mode of the LM2647, the Pulse-skip
(SKIP) Mode. In SKIP Mode, a zero-cross detector at the
SW pin turns off the bottom FET when the inductor current
decays to zero (actually at VSW_ZERO, see Electrical Char-
acteristics table). This would however still amount to conven-
tional DCM, with its attendant problems at extremely light
loads as described earlier. The LM2647 however avoids the
random skipping behavior described earlier, and replaces it
with a more defined or formal SKIP mode. In conventional
DCM, a converter would try to reduce its duty cycle from the
CCM value as the load decreases, as explained previously.
So it would start with the CCM duty cycle value (at the
CCM-DCM boundary), but as the load decreases, the duty
cycle would try to shrink to zero. However, in the LM2647,
the DCM duty cycle is not allowed to fall below 85% of the
CCM value. So when the theoretically required DCM duty
cycle value falls below what the LM2647 is allowed to deliver
(in this mode), pulse-skipping starts. It will be seen that
several of these excess pulses may be delivered, until the
output capacitors charge up enough to notify the error am-
plifier and cause its output to reverse. Thereafter several
pulses could be skipped entirely until the output of the error
amplifier again reverses. The SKIP mode therefore leads to
a reduction in the average switching frequency. Switching
losses and FET driver losses, both of which are proportional
to frequency, are significantly reduced at very light loads and
efficiency is boosted. SKIP mode also reduces the circulat-
ing currents and energy associated with the FPWM mode.
See Figure 3 for a typical plot of SKIP mode at very light
loads. Note the bunching of several fixed-width pulses fol-
lowed by skipped pulses. The average frequency can actu-
ally fall very low at very light loads. Note however that when
this happens the inductor core is seeing only very mild flux
excursions, and so no significant audible noise is created.
But if EMI is a particularly sensitive issue for the particular
application, the user can simply opt for the slightly less
efficient, though constant frequency FPWM mode.
CH1: HDRV, CH2: LDRV, CH3: SW, CH4: IL (0.2A/div)
Output 1V @ 0.04A, VIN = 10V, SKIP, L = 10µH, f = 300kHz
20056311
FIGURE 3. Normal SKIP Mode Operation at Light
Loads
The SKIP mode is enabled when the FPWM pin is held low
(or left floating). Note that at higher loads, and under steady
state conditions (above CCM-DCM boundary), there will be
absolutely no difference in the behavior of the LM2647 or the
associated converter waveforms based on the voltage ap-
plied on the FPWM pin. The differences show up only at light
loads.
Under startup too, since the currents are high until the output
capacitors have charged up, there will be no observable
difference in the shape of the ramp-up of the output rails in
either SKIP mode or FPWM mode. The design has thus
forced the startup waveforms to be identical irrespective of
whether the FPWM mode or the SKIP mode has been
selected.
The designer must realize that even at zero load condition,
there is circulating current when operated in FPWM mode.
This is illustrated in Figure 4. Since duty cycle is the same as
for conventional CCM, from V = L* I / t it can be seen that
I (or Ipp in Figure 4) must remain constant for any load,
including zero. At zero load, the average current through the
inductor is zero, so the geometric center of the sawtooth
waveform (the center being always equal to load current) is
along the x-axis. At critical conduction (boundary between
conventional CCM and what should have been DCM were it
not in FPWM mode), the load current is equal to Ipp/2. Note
that excessively low values of inductance will produce much
higher current ripple and this will lead to higher circulating
currents and dissipation.
11 www.national.com

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