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PDF MC34067 Data sheet ( Hoja de datos )

Número de pieza MC34067
Descripción (MC33067 / MC34067) High Performance Resonant Mode Controllers
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MC34067, MC33067,
NCV33067
High Performance
Resonant Mode Controllers
The MC34067/MC33067 are high performance zero voltage switch
resonant mode controllers designed for off−line and dc−to−dc
converter applications that utilize frequency modulated constant
off−time or constant deadtime control. These integrated circuits
feature a variable frequency oscillator, a precise retriggerable
one−shot timer, temperature compensated reference, high gain wide
bandwidth error amplifier, steering flip−flop, and dual high current
totem pole outputs ideally suited for driving power MOSFETs.
Also included are protective features consisting of a high speed fault
comparator and latch, programmable soft−start circuitry, input
undervoltage lockout with selectable thresholds, and reference
undervoltage lockout. These devices are available in dual−in−line and
surface mount packages.
Features
Zero Voltage Switch Resonant Mode Operation
Variable Frequency Oscillator with a Control Range
Exceeding 1000:1
Precision One−Shot Timer for Controlled Off−Time
Internally Trimmed Bandgap Reference
4.0 MHz Error Amplifier
Dual High Current Totem Pole Outputs
Selectable Undervoltage Lockout Thresholds with Hysteresis
Enable Input
Programmable Soft−Start Circuitry
Low Startup Current for Off−Line Operation
NCV Prefix for Automotive and Other Applications Requiring
Unique Site and Control Change Requirements; AEC−Q100
Qualified and PPAP Capable
These Devices are Pb−Free, Halogen Free/BFR Free and are RoHS
Compliant
15
VCC
Enable / 9
UVLO Adjust
1
OSC Charge
2
OSC RC
Oscillator 3
Control Current
16
One-Shot
Error Amp 6
Output
Noninverting 8
Input
Inverting Input 7
11
Soft-Start
VCC UVLO /
Enable
Variable
Frequency
Oscillator
One-Shot
2.5 V
Clamp
Error
Amp
5.0 V
Reference
Vref UVLO
Steering
Flip-Flop
Soft-Start
Fault Detector/
Latch
5
Vref
14
Output A
12
Output B
13
Pwr GND
10
Fault Input
4 Ground
Figure 1. Simplified Block Diagram
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MARKING
DIAGRAMS
16
1
PDIP−16
P SUFFIX
CASE 648
16
1
MC3x067P
AWLYYWWG
16
1
SOIC−16W
DW SUFFIX
CASE 751G
16
MC3x067DW
AWLYYWWG
1
x = 3 or 4
A = Assembly Location
WL = Wafer Lot
YY = Year
WW = Work Week
G = Pb−Free Package
PIN CONNECTIONS
OSC Charge 1
16 One-Shot RC
OSC RC 2
15 VCC
OSC Control Current 3
14 Drive Output A
GND 4
13 Power GND
Vref 5
12 Drive Output B
Error Amp Out 6
11 CSoft-Start
Inverting Input 7
Noninverting Input 8
10 Fault Input
9
Enable/UVLO
Adjust
(Top View)
ORDERING INFORMATION
See detailed ordering and shipping information in the package
dimensions section on page 2 of this data sheet.
© Semiconductor Components Industries, LLC, 2015
August, 2015 − Rev. 15
1
Publication Order Number:
MC34067/D

1 page




MC34067 pdf
MC34067, MC33067, NCV33067
500
COSC = 200 pF
400
COSC = 300 pF
300 COSC = 500 pF
200
100
0
0
VCC = 12 V
RVFO =
RT =
CT = 500 pF
TA = 25°C
Oscillator Discharge Time is Measured at the Drive Outputs.
20 40 60 80 100
tdischg, OSCILLATOR DISCHARGE TIME (ms)
Figure 2. Oscillator Timing Resistor
versus Discharge Time
3500
3000
2500
2000
1500
1000
500
0
0
VCC = 12 V
TA = 25°C
ROSC = 18.2 k
COSC = 300 pF
400 800 1200 1600
IOSC, OSCILLATOR CONTROL CURRENT (mA)
Figure 3. Oscillator Frequency versus
Oscillator Control Current
2000
0.35
0.30
0.25
0.20
0.15
0.10
0.05
0
0.5 1.0 1.5 2.0 2.5
IOSC, OSCILLATOR CONTROL CURRENT (mA)
3.0
Figure 4. Error Amp Output Low State Voltage
versus Oscillator Control Current
60
VCC = 12 V
COSC = 500 pF
30 ROSC = 100 k
TA = 25°C
20
CT = 300 pF
10
CT = 200 pF
CT = 500 pF
6.0
3.0
0.1
One-Shot Period is Measured
at the Drive Outputs.
0.3 0.6 1.0
3.0
tOS, ONE-SHOT PERIOD (ms)
6.0 10
Figure 5. One−Shot Timing Resistor
versus Period
50 50
VCC = 12 V
40
Gain
VO = 2.0 V
RL = 100 k
60
30
TA = 25°C
70
20 80
10
0
-10
- 20
10 k
Phase
Phase
Margin
= 64°
100 k
1.0M
f, FREQUENCY (Hz)
90
100
110
120
10M
Figure 6. Open Loop Voltage Gain and Phase
versus Frequency
0 *Vref = 5.0 V
- 10
- 20
*Vref = 5.1 V
- 30
- 40
VCC = 12 V
RL =
*Vref at TA = 25°C
- 50
- 55
*Vref = 5.2 V
- 25 0 25 50 75
TA, AMBIENT TEMPERATURE (°C)
100 125
Figure 7. Reference Output Voltage Change
versus Temperature
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MC34067 arduino
MC34067, MC33067, NCV33067
UVLO
UVLO + Fault
Fault
9.0 mA
Error Amp
Clamp
R
Q
S
Fault
Latch
Fault Fault
Comparator Input
10
1.0 V
Soft-Start
CSoft-Start Buffer
11
6 Ground
Figure 20. Fault Detector and Soft−Start
Soft−Start Circuit
The Soft−Start circuit shown in Figure 20 forces the
variable frequency Oscillator to start at the maximum
frequency and ramp downward until regulated by the
feedback control loop. The external capacitor at the
CSoft−Start terminal is initially discharged by the
UVLO+Fault signal. The low voltage on the capacitor
passes through the Soft−Start Buffer to hold the Error
Amplifier output low. After UVLO+Fault switches to a
logic zero, the soft−start capacitor is charged by a 9.0 mA
current source. The buffer allows the Error Amplifier output
to follow the soft−start capacitor until it is regulated by the
Error Amplifier inputs. The soft−start function is generally
applicable to controllers operating below resonance and can
be disabled by simply opening the CSoft−Start terminal.
APPLICATIONS INFORMATION
The MC34067 is specifically designed for zero voltage
switching (ZVS) quasi−resonant converter (QRC)
applications. The IC is optimized for double−ended
push−pull or bridge type converters operating in continuous
conduction mode. Operation of this type of ZVS with
resonant properties is similar to standard push−pull or bridge
circuits in that the energy is transferred during the transistor
on−time. The difference is that a series resonant tank is
usually introduced to shape the voltage across the power
transistor prior to turn−on. The resonant tank in this
topology is not used to deliver energy to the output as is the
case with zero current switch topologies. When the power
transistor is enabled the voltage across it should already be
zero, yielding minimal switching loss. Figure 21 shows a
timing diagram for a half−bridge ZVS QRC. An application
circuit is shown in Figure 22. The circuit built is a dc to dc
half−bridge converter delivering 75 W to the output from a
48 V source.
When building a zero voltage switch (ZVS) circuit, the
objective is to waveshape the power transistor’s voltage
waveform so that the voltage across the transistor is zero
when the device is turned on. The purpose of the control IC
is to allow a resonant tank to waveshape the voltage across
the power transistor while still maintaining regulation. This
is accomplished by maintaining a fixed deadtime and by
varying the frequency; thus the effective duty cycle is
changed.
Primary side resonance can be used with ZVS circuits. In
the application circuit, the elements that make the resonant
tank are the primary leakage inductance of the transformer
(LL) and the average output capacitance (COSS) of a power
MOSFET (CR).
The desired resonant frequency for the application circuit
is calculated by Equation 6:
ƒr =
2π
1
L L 2CR
(eq. 6)
In the application circuit, the operating voltage is low and
the value of COSS versus Drain Voltage is known. Because
the COSS of a MOSFET changes with drain voltage, the
value of the CR is approximated as the average COSS of the
MOSFET. For the application circuit the average COSS can
be calculated by Equation 7:
CR =
2 * COSS measured at
1
2
Vin
(eq. 7)
The MOSFET chosen fixes CR and that LL is adjusted to
achieve the desired resonant frequency.
However, the desired resonant frequency is less critical
than the leakage inductance. Figure 21 shows the primary
current ramping toward its peak value during the resonant
transition. During this time, there is circulating current
flowing through the secondary inductance, which
effectively makes the primary inductance appear shorted.
Therefore, the current through the primary will ramp to its
peak value at a rate controlled by the leakage inductance and
the applied voltage. Energy is not transferred to the
secondary during this stage, because the primary current has
not overcome the circulating current in the secondary. The
larger the leakage inductance, the longer it takes for the
primary current to slew. The practical effect of this is to
lower the duty cycle, thus reducing the operating range.
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