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PDF OPA643 Data sheet ( Hoja de datos )
| Número de pieza | OPA643 | |
| Descripción | Wideband Low Distortion / High Gain OPERATIONAL AMPLIFIER | |
| Fabricantes | Burr-Brown | |
| Logotipo |  |
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Hay una vista previa y un enlace de descarga de OPA643 (archivo pdf) en la parte inferior de esta página. Total 17 Páginas | ||
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OPA643
OOPPAA66453 8
OPA643
Wideband Low Distortion, High Gain
OPERATIONAL AMPLIFIER
FEATURES
q LOW DISTORTION: –90dBc at 5MHz
q LOW NOISE: 2.3nV/√Hz
q GAIN-BANDWIDTH PRODUCT: 800MHz
q AVAILABLE IN SOT23-5 PACKAGE
q STABLE IN GAINS ≥ 3
q HIGH SLEW RATE: 1000V/µs
q HIGH OPEN-LOOP GAIN: 95dB
q HIGH OUTPUT CURRENT: ±60mA
APPLICATIONS
q BASE STATION ADC PREAMP
q ADC/DAC BUFFER AMPLIFIER
q LOW DISTORTION IF AMPLIFIER
q LOW NOISE, BROADBAND,
TRANSIMPEDANCE AMPLIFIER
q LOW NOISE PREAMPLIFIER
q VIDEO AMPLIFICATION
q TEST INSTRUMENTATION
DESCRIPTION
The OPA643 provides a level of speed and dynamic
range previously unattainable in a monolithic op amp.
Using a de-compensated voltage feedback architec-
ture with two internal gain stages, the OPA643 achieves
exceptionally low harmonic distortion over a wide
frequency range. The "classic" differential input pro-
vides all the familiar benefits of precision op amps,
such as bias current cancellation and very low invert-
ing current noise compared with wideband current
feedback op amps. High slew rate and open-loop gain,
along with low input noise and high output current
Low Gain
Compensation
0.1µF
+5V
280Ω
OPA643
2Vp-p 0.1µF
drive make the OPA643 ideal for very high dynamic
range requirements.
The high gain bandwidth product for the gain ≥ 3
stable OPA643 makes it particularly suitable for
wideband transimpedance amplifiers and moderate gain
IF amplifier applications. External compensation
techniques may be used to apply the OPA643 at low
gains giving exceptionally low distortion and frequency
response flatness. Where unity gain stability with
comparable distortion performance is required, consider
the OPA642.
+5V
5kΩ
REFT
0.1µF
50Ω
47pF
ADS805
Analog
Input
12-Bit
20MSPS
Measured
80dB SFDR
1Vp-p
10MHz
50Ω
Source
402Ω
56.9Ω
–5V
806Ω
14pF
2.7pF
5kΩ
REFB
0.1µF
High Dynamic Range 20MSPS Digitizer
International Airport Industrial Park • Mailing Address: PO Box 11400, Tucson, AZ 85734 • Street Address: 6730 S. Tucson Blvd., Tucson, AZ 85706 • Tel: (520) 746-1111 • Twx: 910-952-1111
Internet: http://www.burr-brown.com/ • FAXLine: (800) 548-6133 (US/Canada Only) • Cable: BBRCORP • Telex: 066-6491 • FAX: (520) 889-1510 • Immediate Product Info: (800) 548-6132
©1993 Burr-Brown Corporation
PDS-11191D
OPA643Printed in U.S.A. March, 1998
®
1 page  
TYPICAL PERFORMANCE CURVES (CONT)
At TA = +25°C, VS = ±5V, RL = 100Ω, RF = 402Ω, unless otherwise noted.
5MHz 2ND HARMONIC DISTORTION
–70
G = +5
–75
RL = 200Ω
–80
–85 RL = 100Ω
–90
–95
–100
0.1
RL = 500Ω
1
Output Voltage Swing (Vp-p)
10
5MHz 3RD HARMONIC DISTORTION
–70
G = +5
–75
–80
–85
–90
–95
–100
0.1
RL = 200Ω
RL = 500Ω
1
Output Voltage Swing (Vp-p)
RL = 100Ω
10
10MHz 2ND HARMONIC DISTORTION
–60
G = +5
–65 RL = 100Ω
–70
–75 RL = 200Ω
–80
–85 RL = 500Ω
–90
0.1
1
Output Voltage Swing (Vp-p)
10
10MHz 3RD HARMONIC DISTORTION
–60
G = +5
–65
–70
–75
–80
–85
–90
0.1
RL = 200Ω
RL = 500Ω
1
Output Voltage Swing (Vp-p)
RL = 100Ω
10
20MHz 2ND HARMONIC DISTORTION
–60
G = +5
–65
RL = 100Ω
–70
–75 RL = 200Ω
–80
RL = 500Ω
–85
–90
0.1
1
Output Voltage Swing (Vp-p)
10
20MHz 3RD HARMONIC DISTORTION
–60
G = +5
–65
–70
–75
–80
–85
–90
0.1
RL = 200Ω
RL = 500Ω
RL = 100Ω
1
Output Voltage Swing (Vp-p)
10
®
5 OPA643
5 Page 
This will increase the Q for the closed-loop poles, peaking
up the frequency response and extending the bandwidth. A
peaked frequency response will show overshoot and ringing
in the pulse response as well as a higher integrated output
noise. Operating at a noise gain less than +3 runs the risk of
sustained oscillation (loop instability). However, operation
at low gains would be desirable to take advantage of the
much higher slew rate and lower input noise voltage available
in the OPA643, as compared to performance offered by
unity gain stable op amps. Numerous external compensation
techniques have been suggested for operating a high gain op
amp at low gains. Most of these give zero/pole pairs in the
closed-loop response that cause long term settling tails in the
pulse response and/or phase non-linearity in the frequency
response. Figure 5 shows an external compensation method
for the non-inverting configuration that does not suffer from
these drawbacks.
50Ω Source
+5V
RT
50Ω
RI
133Ω
OPA643
VO 50Ω
RF
402Ω
RG
402Ω
–5V
FIGURE 5. Broadband Low Gain Non-Inverting External
Compensation.
The RI resistor across the two inputs will increase the noise
gain (i.e. decrease the loop gain) without changing the signal
gain. This approach will retain the full slew rate to the output
but will give up some of the low noise benefit of the
OPA643. Assuming a low source impedance, set RI so that
1+RF/(RG || RI) is ≥ +3.
Where a low gain is desired, and inverting operation is
acceptable, a new external compensation technique may be
used to retain the full slew rate and noise benefits of the
OPA643 while maintaining the increased loop gain and the
associated improvement in distortion offered by the
decompensated architecture. This technique shapes the loop
gain for good stability while giving an easily controlled
second-order low pass frequency response. Figure 6 shows
this circuit (the same amplifier circuit as shown on the front
page). Considering only the noise gain for the circuit of
Figure 6, the low frequency noise gain, (NG1) will be set by
the resistor ratios while the high frequency noise gain (NG2)
will be set by the capacitor ratios. The capacitor values set
both the transition frequencies and the high frequency noise
gain. If this noise gain, determined by NG2 = 1+ CS/CF, is set
to a value greater than the recommended minimum stable
gain for the op amp and the noise gain pole, set by 1/RFCF,
is placed correctly, a very well controlled second-order low
pass frequency response will result.
+5V
0.1µF
RT
280Ω
OPA643
RG
402Ω
VI
CS
12.6pF
–5V
RF
806Ω
CF
1.9pF
VO
FIGURE 6. Broadband Low Gain Inverting External
Compensation.
To choose the values for both CS and CF, two parameters and
only three equations need to be solved. The first parameter
is the target high frequency noise gain NG2, which should be
greater than the minimum stable gain for the OPA643. Here,
a target NG2 of 7.5 will be used. The second parameter is
the desired low frequency signal gain, which also sets the
low frequency noise gain NG1. To simplify this discussion,
we will target a maximally flat second-order low pass
Butterworth frequency response (Q = 0.707). The signal
gain of –2 shown in Figure 6 will set the low frequency noise
gain to NG1 = 1 + RF/RG (= 3 in this example). Then, using
only these two gains and the Gain Bandwidth Product (GBP)
for the OPA643 (800MHz), the key frequency in the
compensation can be determined as:
ZO
=
GBP
NG12
1 –
NG1
NG2
–
1–
2
NG1
NG2
Physically, this Z0 (13.6MHz for the values shown in Figure
6) is set by 1/(2π • RF(CF + CS)) and is the frequency at
which the rising portion of the noise gain would intersect
unity gain if projected back to 0dB gain. The actual zero in
the noise gain occurs at NG1 • Z0 and the pole in the noise
gain occurs at NG2 • Z0. Since GBP is expressed in Hz,
multiply Z0 by 2π and use this to get CF by solving:
CF
=
1
2π • RFZONG2
Finally, since CS and CF set the high frequency noise gain,
determine CS by:
CS = (NG2 – 1) CF
The resulting closed-loop bandwidth will be approximately
equal to:
F –3dB ≅ ZO GBP
®
11 OPA643
11 Page | ||
| Páginas | Total 17 Páginas | |
| PDF Descargar | [ Datasheet OPA643.PDF ] | |
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